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rev. 1.2 october 2009 www.aosmd.com page 1 of 14 aoz1013 ezbuck? 3a simple regulator n o t r e c o m m e n d e d f o r n e w d e s i g n s . general description the aoz1013 is a high efficiency, simple to use, 3a buck regulator. the aoz1013 works from a 4.5v to 16v input voltage range, and provides up to 3a of continuous output current with an output voltage adjustable down to 0.8v. the aoz1013 comes in an so-8 package and is rated over a -40c to +85c ambient temperature range. features 4.5v to 16v operating input voltage range 50m ? internal pfet switch for high efficiency: up to 95% internal schottky diode internal soft start output voltage adjustable to 0.8v 3a continuous output current fixed 500khz pwm operation cycle-by-cycle current limit short-circuit protection thermal shutdown small size so-8 package applications point of load dc/dc conversion pcie graphics cards set top boxes dvd drives and hdd lcd panels cable modems telecom/networking/datacom equipment typical application figure 1. 3.3v/3a buck regulator lx vin u1 vout vin from pc fb gnd en comp agnd c1 22f c3 100f r1 r2 10k rs nu r c c c c6 nc c5 1000pf c4 nu cs nu l1 aoz1013 c2 10f d1
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 2 of 14 ordering information * not recommended for new designs. replacement part is aoz1017. all aos products are offering in packaging with pb -free plating and compliant to rohs standards. please visit www.aosmd.com/web/qualit y/rohs_compliant.jsp for additional information. pin configuration pin description block diagram part number ambient temperature range package environmental aoz1013ai* -40c to +85c so-8 rohs lx lx en comp 1 2 3 4 vin pgnd agnd fb so-8 (top view) 8 7 6 5 pin number pin name pin function 1v in supply voltage input. when v in rises above the uvlo thre shold the device starts up. 2 pgnd power ground. electrically needs to be connected to agnd. 3 agnd reference connection for cont roller section. also used as thermal connection for controller section. electrically needs to be connected to pgnd. 4 fb the fb pin is used to determine the output voltage via a resistor divider between the output and gnd. 5 comp external loop compensation pin. 6 en the enable pin is active high. connect en pin to v in if not used. do not leave the en pin floating. 7, 8 lx pwm output connection to inducto r. thermal connection for output stage. 500khz oscillator agnd pgnd vin en fb comp lx lx otp internal +5v ilimit pwm control logic 5v ldo regulator uvlo & por softstart reference & bias 0.8v q1 pwm comp level shifter + fet driver isen eamp + ? + ? + ? +
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 3 of 14 absolute maximum ratings exceeding the absolute maximum ratings may damage the device. note: 1. devices are inherently esd s ensitive, handling precautions are required. human body model rating: 1.5k ? in series with 100pf. recommend operating ratings the device is not guaranteed to operate beyond the maximum operating ratings. note: 2. the value of ja is measured with the device mounted on 1-in 2 fr-4 board with 2oz. copper, in a still air environment with t a = 25c. the value in any given application depends on the user's specific board design. electrical characteristics t a = 25c, v in = v en = 12v, v out = 3.3v unless otherwise specified (3 ) note: 3. specification in bold indicate an ambient temperature range of -40c to +85c. these specifications are guaranteed by design. parameter rating supply voltage (v in )18v lx to agnd -0.7v to v in +0.3v en to agnd -0.3v to v in +0.3v fb to agnd -0.3v to 6v comp to agnd -0.3v to 6v pgnd to agnd -0.3v to +0.3v junction temperature (t j ) +150c storage temperature (t s ) -65c to +150c esd rating (1) 2kv parameter rating supply voltage (v in ) 4.5v to 16v output voltage range 0.8v to v in ambient temperature (t a ) -40c to +85c package thermal resistance so-8 ( ja ) (2 ) 82c/w symbol parameter conditions min. typ. max. units v in supply voltage 4.5 16 v v uvlo input under-voltage lockout threshold v in rising v in falling 4.00 3.70 v i in supply current (quiescent) i out = 0, vfb = 1.2v, v en > 1.2v 23 ma i off shutdown supply current v en = 0v 320 ma v fb feedback voltage 0.782 0.8 0.818 v load regulation 0.5 % line regulation 1% i fb feedback voltage input current 200 na v en en input threshold off threshold on threshold 2.0 0.6 v v hys en input hysteresis 100 mv modulator f o frequency 350 500 600 khz d max maximum duty cycle 100 % d min minimum duty cycle 6% error amplifier voltage gain 500 v / v error amplifier transconductance 200 a / v protection i lim current limit 4 5 a over-temperature shutdown limit t j rising t j falling 145 100 c t ss soft start interval 4ms output stage high-side switch on-resistance v in = 12v v in = 5v 40 65 50 85 m ?
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 4 of 14 typical performance characteristics circuit of figure 1. t a = 25c, v in = v en = 12v, v out = 3.3v unless otherwise specified. light load (dcm) operation full load (ccm) operation startup to full load full load to turnoff 50% to 100% load transient no load to turnoff 1 s/div 1 s/div 1ms/div 1ms/div 100 s/div 1s/div vin ripple 50mv/div vo ripple 0.1v/div vin 5v/div vo 1v/div lin 1a/div lo 2a/div vin 5v/div vo 1v/div lin 1a/div vin 5v/div vo 1v/div lin 1a/div vo ripple 50mv/div il 2a/div vlx 10v/div vin ripple 0.1v/div vo ripple 50mv/div il 2a/div vlx 10v/div
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 5 of 14 typical performance characteristics (continued) circuit of figure 1. t a = 25c, v in = v en = 12v, v out = 3.3v unless otherwise specified. short circuit protection short circuit recovery 100 s/div 1ms/div vo 2v/div il 2a/div vo 2v/div il 2a/div aoz1013ai efficiency efficiency (v in = 12v) vs. load current derating curve at 5v input 3.5 3.0 2.5 2.0 1.5 1.0 0 3.3v output 1.8v, 3.3v, 5.0v output 1.8v, 5.0v, 8.0v output 3.3v output 5.0v output 8.0v output 0 0.5 1.0 1.5 2.0 2.5 3.0 25 35 45 55 65 75 85 load current (a) ambient temperature (t a ) efficieny (%) output current (i o ) 3.5 3.0 2.5 2.0 1.5 1.0 0 output current (i o ) derating curve at 12v input 25 35 45 55 65 75 85 ambient temperature (t a ) thermal de-rating curves for so-8 package part under typical input and output condition based on the evaluation board. 25 c ambient temperature and natural convection (air speed < 50lfm) unless otherwise specified. 75 80 85 90 95
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 6 of 14 detailed description the aoz1013 is a current-mode step down regulator with integrated high side pmos switch. it operates from a 4.5v to 16v input voltage range and supplies up to 3a of load current. the duty cycle can be adjusted from 6% to 100% allowing a wide range of output voltage. features include enable control, power-on reset, input under voltage lockout, fixed internal soft-start and thermal shut down. the aoz1013 is available in so-8 package. enable and soft start the aoz1013 has internal soft start feature to limit in-rush current and ensure the output voltage ramps up smoothly to regulation voltage. a soft start process begins when the input voltage rises to 4.0v and voltage on en pin is high. in soft start process, the output voltage is ramped to regulation voltage in typically 4ms. the 4ms soft start time is set internally. the en pin of the aoz1013 is active high. connect the en pin to v in if enable function is not used. pulling en to ground will disable the aoz1013 . do not leave it open. the voltage on en pin must be above 2.0v to enable the aoz1013. when voltage on en pin falls below 0.6v, the aoz1013 is disabled. if an app lication circuit requires the aoz1013 to be disabled, an open drain or open collector circuit should be used to interface to en pin. steady-state operation under steady-state conditions, the converter operates in fixed frequency and continuous-conduction mode (ccm). the aoz1013 integrates an internal p-mosfet as the high-side switch. inductor current is sensed by amplifying the voltage drop across the drain to source of the high side power mosfet. output voltage is divided down by the external voltage divider at the fb pin. the difference of the fb pin voltage and reference is amplified by the internal transconductance error amplifier. the error volt- age, which shows on the comp pin, is compared against the current signal, which is sum of inductor current signal and ramp compensation signal, at pwm comparator input. if the current signal is less than the error voltage, the internal high-side switch is on. the inductor current flows from the input through the inductor to the output. when the current signal exceeds the error voltage, the high-side switch is off. the inductor current is free- wheeling through the external schottky diode to output. the aoz1013 uses a p-channel mosfet as the high side switch. it saves the bootstrap capacitor normally seen in a circuit which is using an nmos switch. it allows 100% turn-on of the upper switch to achieve linear regu- lation mode of operation. the minimum voltage drop from v in to v o is the load current times dc resistance of mosfet plus dc resistance of buck inductor. it can be calculated by equation below: where; v o_max is the maximum output voltage, v in is the input voltage from 4.5v to 16v, i o is the output current from 0a to 3a, r ds(on) is the on resistance of in ternal mosfet, the value is between 40m ? and 70m ? depending on input voltage and junction temperature, and r inductor is the inductor dc resistance. switching frequency the aoz1013 switching frequency is fixed and set by an internal oscillator. the pr actical switching frequency could range from 350khz to 600khz due to device variation. output voltag e programming output voltage can be set by feeding back the output to the fb pin with a resistor divider network. in the application circuit shown in figure 1. the resistor divider network includes r 1 and r 2 . usually, a design is started by picking a fixed r 2 value and calculating the required r 1 with equation below: some standard values of r 1 and r 2 for the most com- monly used output voltage values are listed in table 1. table 1. v o (v) r 1 (k ? ) r 2 (k ? ) 0.8 1.0 open 1.2 4.99 10 1.5 10 11.5 1.8 12.7 10.2 2.5 21.5 10 3.3 31.6 10 5.0 52.3 10 v o_max v in i o r ds on () r inductor + () ? = v o 0.8 1 r 1 r 2 ------ - + ?? ?? ?? =
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 7 of 14 the combination of r 1 and r 2 should be large enough to avoid drawing excessive current from the output, which will cause power loss. since the switch duty cycle can be as high as 100%, the maximum output voltage can be set as high as the input voltage minus the voltage drop on upper pmos and inductor. protection features the aoz1013 has multiple prot ection features to prevent system circuit damage unde r abnormal conditions. over current protection (ocp) the sensed inductor current signal is also used for over current protection. since the aoz1013 employs peak current mode control, the comp pin voltage is proportional to the peak inductor current. the comp pin voltage is limited to be between 0.4v and 2.5v internally. the peak inductor current is automatically limited cycle by cycle. the cycle by cycle current limit threshold is set between 4a and 5a. when the load current reaches the current limit threshold, the cycle by cy cle current limit circuit turns off the high side switch immediately to terminate the current duty cycle. the inductor current stops rising. the cycle by cycle current limit protection directly limits inductor peak current. the average inductor current is also limited due to the limitation on peak inductor current. when the cycle by cycle current limit circuit is triggered, the output voltage drops as the duty cycle is decreasing. the aoz1013 has internal short circuit protection to protect itself from catastrophic failure under output short circuit conditions. the fb pin voltage is proportional to the output voltage. whenever fb pin voltage is below 0.2v, the short circuit protection circuit is triggered. as a result, the converter is shut down and hiccups at a frequency equal to 1/8 of normal switching frequency. the converter will start up via a soft start once the short circuit condition disappears. in short circuit protection mode, the inductor average current is greatly reduced because of the low hiccup frequency. power-on reset (por) a power-on reset circuit monitors the input voltage. when the input voltage exceeds 4v, the converter starts operation. when input voltage falls below 3.7v, the converter shuts down. thermal protection an internal temperature sensor monitors the junction temperature. it shuts down the internal control circuit and high side pmos if the junction temperature exceeds 145c. the regulator will rest art automatica lly under the control of soft-start circuit when the junction temperature decreases to 100c. application information the basic aoz1013 application circuit is shown in figure 1. component selection is explained below. input capacitor the input capacitor must be connected to the vin pin and pgnd pin of the aoz1013 to maintain steady input voltage and filter out the puls ing input current. the volt- age rating of input capacitor must be greater than maxi- mum input voltage plus ripple voltage. the input ripple voltage can be approximated by the equation below: since the input current is discontinuous in a buck converter, the current stress on the input capacitor is another concern when selecting the capacitor. for a buck circuit, the rms value of input capacitor current can be calculated by: if let m equal the conversion ratio: the relationship between the input capacitor rms cur- rent and voltage conversion ratio is calculated and shown in figure 2 on the next page. it can be seen that when v o is half of v in , c in is under the worst current stress. the worst current stress on c in is 0.5 x i o . v in i o fc in ----------------- 1 v o v in -------- - ? ?? ?? ?? v o v in -------- - = i cin_rms i o v o v in -------- - 1 v o v in -------- - ? ?? ?? ?? = v o v in -------- - m =
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 8 of 14 figure 2. i cin vs. voltage conversion ratio for reliable operation and best performance, the input capacitors must have current rating higher than i cin_rms at the worst operating conditions. ceramic capacitors are preferred for input capacitors because of their low esr and high ripple current rating. depending on the application circuits, other low esr tantalum capacitors or aluminum electrolytic capacitors may also be used. when selecting ceramic capacitors, x5r or x7r type dielectric ceramic capacitors ar e preferred for their better temperature and voltage characteristics. note that the ripple current rating from capacitor manufactures is based on certain amount of life time. further de-rating may be necessary for practical design requirement. inductor the inductor is used to supp ly constant current to the output when it is driven by a switching voltage. for a given input and output voltage, inductance and switching frequency together decide the inductor ripple current, which is: the peak inductor current is: high inductance gives low indu ctor ripple current but requires a larger size inductor to avoid saturation. low ripple current reduces inductor core losses. low ripple current also reduces rms current through the inductor and switches, which results in less conduction loss. when selecting the inductor, make sure it is able to handle the peak current at the highest operating temper- ature without saturation. the inductor takes the highest current in a buck circuit. the conduction loss on the inductor needs to be checked for thermal and efficiency requirements. surface mount inductors in different shape and styles are available from coilcraft, elytone and murata. shielded inductors are small and radiate less emi noise, but they cost more than unshielded inductors. the choice depends on emi requirement, price and size. table 2 lists some inductors for typical output voltage design. output capacitor the output capacitor is selected based on the dc output voltage rating, output ripple voltage specification, and ripple current rating. the selected output capacit or must have a higher rated voltage specification than the maximum desired output voltage including ripple. de-rating needs to be considered for long term reliability. output ripple voltage specif ication is another important factor for selecting the output capacitor. in a buck converter circuit, output ripple voltage is determined by inductor value, switching fr equency, output capacitor 0 0.1 0.2 0.3 0.4 0.5 0 0.5 1 m i cin_rms (m) i o i l v o fl ---------- - 1 v o v in -------- - ? ?? ?? ?? = i lpeak i o i l 2 -------- + = table 2. typical inductors v out l1 manufacture 5.0v shielded, 6.8h, mss1278-682mld coilcraft shielded, 6.8h, mss1260-682mld coilcraft 3.3v un-shielded, 4.7h, do3316p-472mld coilcraft shielded, 4.7h, do1260-472nxd coilcraft shielded, 3.3h, et553-3r3 elytone 1.8v shield, 2.2h, et553-2r2 elytone un-shielded, 3.3h, do3316p-222mld coilcraft shielded, 2.2h, mss1260-222nxd coilcraft
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 9 of 14 value and esr. it can be calculated by the equation below: where, c o is output capacitor value, and esr co is the equivalent series resistance of the output capacitor. when low esr ceramic capacitor is used as output capacitor, the impedance of the capacitor at the switch- ing frequency dominates. output ripple is mainly caused by capacitor value and inductor ripple current. the output ripple voltage calculation can be simplified to: if the impedance of esr at switching frequency dominates, the output ripple voltage is mainly decided by capacitor esr and inductor ripple current. the output ripple voltage calculation can be further simplified to: for lower output ripple voltage across the entire operat- ing temperature range, x5r or x7r dielectric type of ceramic, or other low esr tantalum capacitor or alumi- num electrolytic capacitor may also be used as output capacitors. in a buck converter, output capacitor current is continuous. the rms current of output capacitor is decided by the peak to peak inductor ripple current. it can be calculated by: usually, the ripple cu rrent rating of the output capacitor is a smaller issue because of the low current stress. when the buck inductor is selected to be very small and inductor ripple current is high, output capacitor could be overstressed. loop compensation the aoz1013 employs peak current mode control for easy use and fast transient response. peak current mode control eliminates the doubl e pole effect of the output l&c filter. it greatly simp lifies the compensation loop design. with peak current mode control, the buck power stage can be simplified to be a one-pole and one-zero system in frequency domain. the pole is dominant pole and can be calculated by: the zero is a esr zero due to output capacitor and its esr. it is can be calculated by: where; c o is the output filter capacitor, r l is load resistor value, and esr co is the equivalent series resi stance of output capacitor. the compensation design is actually to shape the converter close loop transfer function to get desired gain and phase. several different types of compensation networks can be used for aoz1013. for most cases, a series capacitor and resistor network connected to the comp pin sets the pole-zero and is adequate for a stable high-bandwidth control loop. in the aoz1013, fb pin and comp pin are the inverting input and the output of internal transconductance error amplifier. a series r and c compensation network connected to comp provides one pole and one zero. the pole is: where; g ea is the error amplifier transconductance, which is 200 x 10 -6 a/v, g vea is the error amplifier voltage gain, which is 500 v/v, and c c is compensati on capacitor. the zero given by the external compensation network, capacitor c c and resistor r c ,is located at: to design the compensation circuit, a target crossover frequency f c for close loop must be selected. the system crossover frequency is where control loop has unity gain. the crossover frequency is also called the converter bandwidth. generally a higher bandwidth means faster response to load transient. however, the bandwidth should not be too high because of system stability v o i l esr co 1 8 fc o ------------------------- + ?? ?? = v o i l 1 8 fc o ------------------------- = v o i l esr co = i co_rms i l 12 ---------- = f p 1 1 2 c o r l ---------------------------------- - = f z 1 1 2 c o esr co ------------------------------------------------ = f p 2 g ea 2 c c g vea ------------------------------------------ - = f z 2 1 2 c c r c ----------------------------------- =
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 10 of 14 concerns. when designing the compensation loop, converter stability under all li ne and load condition must be considered. usually, it is recommended to set the bandwidth to be less than 1/10 of switching frequency. the aoz1013 operates at a fixed switching frequency range from 350khz to 600khz. the recommended crossover frequency is less than 30khz. the strategy for choosing r c and c c is to set the cross over frequency with r c and set the compensator zero with c c . using selected cr ossover frequency, f c , to calculate r c : where; f c is the desired crossover frequency, v fb is 0.8v, g ea is the error amplifier transco nductance, which is 200 x 10 -6 a/v, and g cs is the current sense circui t transconductance, which is 6.68 a/v. the compensation capacitor c c and resistor r c together make a zero. this zero is put somewhere close to the dominate pole, f p1 , but lower than 1/5 of the selected crossover frequency. c c can is selected by: the previous equation can also be simplified to: an easy-to-use application software which helps to design and simulate the compensation loop can be found at www.aosmd.com . table 3 lists the values for a typical output voltage design when output is 44f ceramics capacitor. table 3. thermal management and layout consideration in the aoz1013 buck regulator circuit, high pulsing cur- rent flows through two circuit loops. the first loop starts from the input capacitors, to the v in pin, to the lx pins, to the filter inductor, to the ou tput capacitor and load, and then return to the input capacitor through ground. current flows in the first loop when t he high side switch is on. the second loop starts from inductor, to the output capacitors and load, to the anode of schottky diode, to the cathode of schottky diode. current flows in the second loop when the low side diode is on. in pcb layout, minimizing the two loops area reduces the noise of this circuit and im proves efficiency. a ground plane is strongly recommended to connect input capaci- tor, output capacitor, and pgnd pin of the aoz1013. in the aoz1013 buck regulator circuit, the two major power dissipating components are the aoz1013, the schottky diode, and output inductor. the total power dissipation of converter circuit can be measured by input power minus output power. the power dissipation in schottky can be approximately calculated as: where; v fw_schottky is the schottky diode forward voltage drop. the power dissipation of in ductor can be approximately calculated by output current and dcr of inductor. the actual junction temperature can be calculated with power dissipation in the aoz1013 and thermal impedance from junction to ambient is: f c 30 khz = r c f c v o v fb ---------- 2 c o g ea g cs ----------------------------- - = c c 1.5 2 r c f p 1 ----------------------------------- = c c c o r l r c --------------------- = v out l1 r c c c 1.8v 2.2h 49.9k ? 1.5nf 3.3v 4.7h 20k ? 2.2nf 5v 6.8h 49.9k ? 1.2nf 8v 10h 49.9k ? 1.2nf p total_loss v in i in v o i o ? = p diode_loss i o 1 d ? () v fw_schottky = p inductor_loss i o 2 r inductor 1.1 = t junction p total_loss p inductor_loss ? () ja =
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 11 of 14 the maximum junction tem perature of aoz1013 is 145c, which limits the maximu m load current capability. please see the thermal de-r ating curves for maximum load current of the aoz1013 under different ambient temperatures. the thermal performance of the aoz1013 is strongly affected by the pcb layout. extra care should be taken by users during the design process to ensure that the ic will operate under the reco mmended environmental conditions. several layout tips are liste d below for the best electric and thermal perfo rmance. figure 3 below illustrates a pcb layout example as a reference. 1. do not use thermal relief connection to the v in and the pgnd pin. pour a maximized copper area to the pgnd pin and the v in pin to help thermal dissipation. 2. input capacitor should be connected to the v in pin and the pgnd pin as close as possible. 3. a ground plane is preferred. if a ground plane is not used, separate pgnd from agnd and connect them only at one point to avoid the pgnd pin noise coupling to the agnd pin. 4. make the current trace from lx pins to l to co to the pgnd as short as possible. 5. pour copper plane on all unused board area and connect it to stable dc nodes, like v in , gnd, or v out . 6. the two lx pins are connected to the internal pfet drain. they are low resistance thermal conduction path and most noisy switching node. connect a copper plane to the lx pin to help thermal dissipation. this copper plane should not be too large otherwise switching noise may be coupled to other parts of the circuit. 7. keep sensitive signal traces away from the lx pins. figure 3. aoz1013 pcb layout so-8 1 2 3 4 8 7 6 5 pgnd agnd comp lx lx en fb cout cc rc cin l vin
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 12 of 14 package dimensions, so-8l notes: 1. all dimensions are in millimeters. 2. dimensions are inclusive of plating 3. package body sizes exclude mold flash and gate burrs. mold flash at the non-lead sides should be less than 6 mils. 4. dimension l is measured in gauge plane. 5. controlling dimension is millimeter, converted inch dimensions are not necessarily exact. symbols a a1 a2 b c d e1 e e h l dimensions in millimeters min. 1.35 0.10 1.25 0.31 0.17 4.80 3.80 5.80 0.25 0.40 0 d c l h x 45 7 (4x) b 2.20 5.74 0.80 unit: mm 1.27 a1 a2 a 0.1 gauge plane seating plane 0.25 e 8 1 e1 e nom. 1.65 ? 1.50 ? ? 4.90 3.90 1.27 bsc 6.00 ? ? ? max. 1.75 0.25 1.65 0.51 0.25 5.00 4.00 6.20 0.50 1.27 8 symbols a a1 a2 b c d e1 e e h l dimensions in inches min. 0.053 0.004 0.049 0.012 0.007 0.189 0.150 0.228 0.010 0.016 0 nom. 0.065 ? 0.059 ? ? 0.193 0.154 0.050 bsc 0.236 ? ? ? max. 0.069 0.010 0.065 0.020 0.010 0.197 0.157 0.244 0.020 0.050 8
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 13 of 14 tape and reel dimensions so-8 carrier tape so-8 reel so-8 tape leader/trailer & orientation tape size 12mm reel size ?330 m ?330.00 0.50 package so-8 (12mm) a0 6.40 0.10 b0 5.20 0.10 k0 2.10 0.10 d0 1.60 0.10 d1 1.50 0.10 e 12.00 0.10 e1 1.75 0.10 e2 5.50 0.10 p0 8.00 0.10 p1 4.00 0.10 p2 2.00 0.10 t 0.25 0.10 n ?97.00 0.10 k0 unit: mm b0 g m w1 s k h n w v r trailer tape 300mm min. or 75 empty pockets components tape orientation in pocket leader tape 500mm min. or 125 empty pockets a0 p1 p2 see note 5 see note 3 see note 3 feeding direction p0 e2 e1 e d0 t d1 w 13.00 0.30 w1 17.40 1.00 h ?13.00 +0.50/-0.20 k 10.60 s 2.00 0.50 g ? r ? v ?
aoz1013 rev. 1.2 october 2009 www.aosmd.com page 14 of 14 aoz1013 package marking z1013ai fay part number code assembly lot code fab & assembly location year & week code wlt as used herein: 1. life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body or (b) support or sustain life, and (c) whose failure to perform when properly used in accordance with instructions for use provid ed in the labeling, can be reasonably expected to result in a significant injury of the user. 2. a critical component in any component of a life support, device, or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. this data sheet contains preliminary data; supplementary data may be published at a later date. alpha & omega semiconductor reserves the right to make changes at any time without notice. life support policy alpha & omega semiconductor products ar e not authorized for use as critical components in life supp ort devices or systems.

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